Systems and methods for recovery of a sub-carrier signal from a stereophonic multiplexed signal

ABSTRACT

Disclosed herein are systems and methods for recovering a sub-carrier signal from a multiplexed signal having an embedded pilot tone signal. The recovery system includes circuitry for recovering a pilot signal from the received multiplexed signal, for generating a frequency-doubled signal from the recovered pilot signal, and for phase-shifting the frequency-doubled signal by a pre-determined phase difference from the embedded pilot tone signal. Another recovery system includes circuitry for recovering a pilot signal from the received multiplexed signal, for phase-shifting the pilot signal by a pre-determined phase difference from the embedded pilot tone signal, and for generating a frequency-doubled signal from the phase-shifted signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.11/582,735, filed Oct. 17, 2006, which claims the benefit of U.S.provisional patent application Ser. No. 60/782,945, filed Mar. 16, 2006,which is hereby incorporated by reference herein in its entirety.

BACKGROUND OF THE INVENTION

Stereo FM receivers are commonly used in many consumer products. In aconventional stereo FM receiver, an input analog signal is received,down-converted to an intermediate frequency (IF), and digitized beforebeing demodulated to a stereophonic multiplexed (MPX) signal. Theresulting MPX signal is composed of three components: (a) a summationsignal of left (L) and right (R) channels

$\left( \frac{L + R}{2} \right)$at baseband, (b) a pilot tone signal at 19 kHz, and (c) adouble-sideband suppressed-carrier (DSBSC) modulated difference signal

$\left( \frac{L - R}{2} \right)$at 38 kHz, twice the frequency of the pilot tone signal.

In an FM stereo receiver, the difference component

$\left( \frac{L - R}{2} \right)$is extracted from the MPX signal and down-converted to baseband beforebeing added to and subtracted from the baseband summation component

$\left( \frac{L + R}{2} \right)$to produce distinct L and R signals for stereo output. Typically,minimal signal processing is required to extract the baseband summationcomponent of the MPX signal. The 19 kHz pilot signal may be recovered bypassing the MPX signal through a narrow band-pass filter centered at 19kHz. This received pilot signal may then undergo frequency doubling togenerate a 38 kHz sub-carrier signal. The difference component may berecovered by first passing the MPX signal through a band-pass filtercentered at 38 kHz. The resulting signal may then be modulated with the38 kHz sub-carrier to down-convert the difference component to baseband.The recovered

$\left( \frac{L - R}{2} \right)$difference component and

$\left( \frac{L + R}{2} \right)$summation component, both in baseband, may then be added and subtractedto generate separate L and R outputs.

In doubling the 19 kHz recovered pilot signal to produce a 38 KHzsub-carrier signal, a strict phase relation between the two signals mustbe enforced to maintain a separation between the left and right channelsin an effort to minimize channel leakage. This is important becausechannel leakage may have a detrimental effect on the resulting audioqualities. Moreover, the non-ideal nature of the circuit components usedin most signal-processing circuitry complicates this phase relationbetween the pilot tone and sub-carrier signals by injecting noise, suchas phase delays, into the waveforms.

This invention relates to systems and methods for producing a 38 kHzsub-carrier signal from a 19 kHz pilot tone signal and for maintaining aphase relation between the two signals in an effort to minimize channelleakage.

SUMMARY OF THE INVENTION

The invention provides systems and methods for recovering a sub-carriersignal from a multiplexed signal having an embedded pilot tone signaland for enforcing a pre-determined phase difference between thesub-carrier and pilot tone signals.

According to one aspect of the invention, a receiver system is providedthat includes circuitry for receiving an input MPX signal having anembedded pilot tone signal. A pilot signal may be recovered from thereceived MPX signal using, for example, a band-pass filter. Afrequency-doubled signal may then be generated from the recovered pilotsignal. The resulting frequency-doubled signal may subsequently be phaseshifted by a pre-determined phase difference from the embedded pilottone signal.

In one aspect, the recovered pilot signal may include a phase differencefrom the pilot tone signal.

In one aspect, the circuitry for generating the frequency-doubled signalfrom the recovered pilot signal may include circuitry for squaring therecovered pilot signal to generate an intermediate signal, circuitry forextracting a high-frequency component of the intermediate signal,circuitry for providing an amplitude scaling function, and circuitry formultiplying the amplitude scaling function with the high-frequencycomponent to generate the frequency-doubled signal.

In one aspect, the circuitry for providing the amplitude scalingfunction includes circuitry for delaying the intermediate signal togenerate a delayed signal, circuitry for subtracting the high-frequencycomponent from the delayed signal to generate a modulating function, andcircuitry for inverting the modulating function to generate theamplitude scaling function.

In one aspect, the circuitry for generating the frequency-doubled signalfrom the recovered pilot signal may include circuitry for providing anestimated derivative of the recovered pilot signal, circuitry formultiplying the recovered pilot signal with the estimated derivative togenerate an intermediate signal, and circuitry for multiplying theintermediate signal with an amplitude-scaling function to generate thefrequency-doubling signal.

In one aspect, the pre-determined phase difference between the embeddedpilot tone signal and the phase-shifted signal is approximately aquadrature phase.

In one aspect, the pre-determined phase difference between the embeddedpilot tone signal and the phase-shifted signal is approximately zero.

According to another aspect of the invention, a receiver system isprovided for recovering a sub-carrier signal from a multiplexed signalhaving an embedded pilot tone signal. The system includes a receiver forreceiving the multiplexed signal. A pilot signal may be recovered fromthe received multiplexed signal using circuitry such as a band-passfilter. The recovered pilot signal may then be phase-shifted by apre-determined phase difference from the embedded pilot tone signal.This pre-determined phase difference may be approximately

$\frac{\pi}{4}.$The phase-shifted signal may then be used to generate afrequency-doubled signal from the phase-shifted signal.

In another aspect, circuitry used for generating the frequency-doubledsignal from the phase-shifted signal includes circuitry for squaring thephase-shifted signal to generate an intermediate signal, circuitry forextracting a high-frequency component of the intermediate signal,circuitry for providing an amplitude scaling function, and circuitry formultiplying the amplitude scaling function with the high-frequencycomponent to generate the frequency-doubled signal.

In another aspect, circuitry used for providing the amplitude scalingfunction includes circuitry for delaying the intermediate signal togenerate a delayed signal, circuitry for subtracting the high-frequencycomponent from the delayed signal to generate a modulating function, andcircuitry for inverting the modulating function to generate theamplitude scaling function.

Further features of the invention, its nature and various advantages,will be more apparent from the accompanying drawings and the followingdetailed description of the preferred embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is an illustrative block diagram of an exemplary 38 kHzsub-carrier signal recovery system in accordance with an aspect of theinvention.

FIG. 2 is an illustrative block diagram of a Frequency-Doubling CosineSignal Normalization (FDSN-COS) circuitry in accordance with an aspectof the invention.

FIG. 3 is an illustrative digital-domain implementation of the FDSN-COScircuitry of FIG. 2.

FIG. 4 depicts an illustrate example of a 38 kHz sub-carrier signalrecovery using the scheme shown in FIG. 1.

FIG. 5 is an illustrative block diagram of another exemplary 38 kHzsub-carrier signal recovery implementation in accordance with an aspectof the invention.

FIG. 6 depicts an illustrate example of a 38 kHz sub-carrier signalrecovery using the scheme shown in FIG. 6.

FIG. 7 is an illustrative block diagram of yet another exemplary 38 kHzsub-carrier signal recovery implementation in accordance with an aspectof the invention.

FIG. 8 is an illustrative block diagram of a Frequency-Doubling SineSignal Normalization (FDSN-SIN) circuitry in accordance with an aspectof the invention.

FIG. 9 is an illustrative digital-domain implementation of the FDSN-SINcircuitry of FIG. 9.

FIG. 10 is an illustrative digital-domain implementation of a phaselook-ahead scheme in accordance with an aspect of the invention.

FIG. 11A is a block diagram of an exemplary high definition televisionthat can employ the disclosed technology.

FIG. 11B is a block diagram of an exemplary vehicle that can employ thedisclosed technology.

FIG. 11C is a block diagram of an exemplary cell phone that can employthe disclosed technology.

FIG. 11D is a block diagram of an exemplary set top box that can employthe disclosed technology.

FIG. 11E is a block diagram of an exemplary media player that can employthe disclosed technology.

DETAILED DESCRIPTION OF THE INVENTION

The invention provides systems and methods for recovering a 38 kHzsub-carrier signal from a 19 kHz pilot tone signal and for enforcing apre-determined phase relation between those signals. For ease ofexplanation and without limiting the scope of the invention, it will beassumed that the sub-carrier and pilot signals are both expressed assinusoidal functions.

FIG. 1 illustrates a high-level block diagram 100 of an exemplary 38 kHzsub-carrier signal recovery scheme according to one aspect of theinvention. In this implementation, the pilot and sub-carrier signals arerepresented as cosine functions. The circuitry includes a 19 kHzband-pass filter 102 for recovering the pilot tone signal, a FrequencyDoubling Scale Normalization (FDSN-COS) functional block 104 forgenerating a 38 kHz sub-carrier signal from the recovered pilot signal,and a phase look-ahead functional block 106 for imposing apre-determined phase difference between the pilot and sub-carriersignals.

The received MPX signal may be expressed as a composite of the followingcosine functions:

$\begin{matrix}{{{MPX} = {{0.9*M^{\prime}} + {0.9*S^{\prime}*{\cos\left( {{2\pi\; f_{38{kHz}}t} + \frac{\pi}{2}} \right)}} + {\left( {0.09 \pm 0.01} \right)*{\cos\left( {2\pi\; f_{19{kHz}}t} \right)}}}},} & {{Equation}\mspace{14mu}(1)}\end{matrix}$where the first term of Equation (1) corresponds to a baseband summationcomponent signal, the second term corresponds to an amplitude-modulateddifference component signal at 38 kHz, and the third term corresponds toa reference pilot tone signal at 19 kHz. In particular, the M′ signal ofEquation (1) represents a pre-emphasis filtered version of the summationcomponent

$\left( \frac{L + R}{2} \right),$and the S′ signal represents a pre-emphasis filtered version of thedifference component

$\left( \frac{L - R}{2} \right).$In general, pre-emphasis filtering (not shown) reduces high-frequencynoise associated with signal transmissions. It should be understood thatany suitable signal processing may be performed with the received MPXsignal to prepare the signal for demodulation. However, in someembodiments, no additional processing is required.

In an effort to maintain a separation between the L and R signals forminimizing potential occurrences of channel leakage, a

$\frac{\pi}{2}$or quadrature phase difference may be introduced between the cosinesub-carrier signal of the second term in Equation (1) and the recoveredcosine pilot signal of the third term. It can be shown that introducingthis quadrature phase separation may ensure the condition that, when Lis a positive value and R=−L, whenever the MPX signal crosses the timeaxis the MPX signal has a positive slope correlating to the occurrenceof pilot tone signal having an instantaneous value of zero. Thisrequirement, herein referred to as a “positive-slope requirement,” maybe maintained by introducing a quadrature phase difference between thepilot and sub-carrier components of the MPX signal of Equation (1).However, the phase condition required to maintain the positive-sloperequirement varies depending on how a MPX signal is represented. Forexample, the quadrature phase offset derived for the cosine MPX signalin Equation (1) would not be the same in the case where the MPX signalis represented as a sine function. This concept will be explained belowin greater detail.

A sharp band-pass filter 102 centered at 19 kHz may then be used torecover the 19 kHz pilot signal from the MPX signal. Ideally, therecovered pilot signal may be represented as a(t)cos(2πf_(19kHz)t),where a(t) is a slow-varying amplitude scaling function, but due to thenon-ideal nature of band-pass filter 102, an unwanted delay may beinjected into the recovered pilot signal at the output of the filteringprocess. The pilot signal including this delay may be represented asa(t)cos(2πf_(19kHz)(t−τ_(bpf))), where τ_(bpf) represents a non-idealdelay introduced by band-pass filter 102. τ_(bpf) may also representother delays introduced by system 100 to the recovered pilot signal.

The recovered pilot signal may then be applied to FDSN-COS functionalblock 104, subsequently generating a normalized 38 kHz signalcos(2πf_(38kHz)(t−τ_(bpf))). It is observed that filter delay τ_(bpf) ispropagated from the recovered pilot signal to the output of FDSN-COSblock 104. Hence phase look-ahead block 106 may then be used to apply aphase correction to the signal output to approximately cancel filterdelay τ_(bpf) while introducing an overall quadrature phase offset tothe normalized 38 kHz signal. FDSN-COS block 104 and quadrature phaselook-ahead block 106 will be described below in further operationaldetail. The resulting signal produced from the arrangement of FIG. 1 isa 38 kHz sub-carrier signal

$\cos\left( {{2\;\pi\; f_{{38\mspace{11mu}{kHz}}\;}t} + \frac{\pi}{2}} \right)$which satisfies the phase condition set forth by the positive-sloperequirement described above.

An exemplary implementation of FDSN-COS functional block 104 isillustrated in FIG. 2. FDSN-COS block 200 operates by producing anormalized 38 kHz signal from an input 19 kHz recovered pilot signal.First a square operation 206 is applied to the input recovered pilotsignal a(t)cos(2πf_(19kHz)(t−τ_(bpf))) to produce a frequency-doubledsignal

$\frac{a^{2}(t)}{2}\left\lbrack {1 + {\cos\left( {2\;\pi\;{f_{38\mspace{11mu}{kHz}}\left( {t - \tau_{bpf}} \right)}} \right)}} \right\rbrack$based on the trigonometric identity

${{\cos^{2}\;\theta} = {\frac{1}{2}\left( {1 + {\cos\; 2\;\theta}} \right)}},$where here θ=2πf_(19kHz)(t−τ_(bpf)). This signal may then be processedby a high-pass filter 208 to remove the baseband component

$\frac{a^{2}(t)}{2}$and is then multiplied by a normalizing function

$\frac{2}{a^{2}(t)}$at multiplier 210 to further produce a normalized signalcos(2πf_(38kHz)(t−τ_(bpf))). This signal preferably has a 38 kHzsampling rate and is in phase with the recovered pilot signal. Thescaling function may be determined by delaying the signal

$\frac{a^{2}(t)}{2}\left\lbrack {1 + {\cos\left( {2\;\pi\;{f_{38\mspace{11mu}{kHz}}\left( {t - \tau_{bpf}} \right)}} \right)}} \right\rbrack$to extract its slow-varying component

$\frac{a^{2}(t)}{2}$before inverting the component to generate the scaling function

$\frac{2}{a^{2}(t)}.$

FIG. 3 provides an exemplary digital-domain implementation 300 ofFDSN-COS block 104. In particular, a digital 19 kHz recovered pilotsignal

$a_{n}{\cos\left( {2\;\pi\; f_{19\mspace{11mu}{kHz}}\;\left( {\frac{n}{f_{s}} - \tau_{bpf}} \right)} \right)}$may be applied as an input to system 300, where n is an integer, f_(s)is a sampling frequency, and τ_(bpf) is a digital phase delay introducedby the use of a band-pass filter (not shown) to recover the pilot tonesignal from a received MPX signal. This recovered pilot signal thenundergoes a square operation 302 to generate a 38 kHz intermediatewaveform

${\frac{a_{n}^{2}}{2}\left\lbrack {1 + {\cos\left( {2\;\pi\;{f_{38\mspace{11mu}{kHz}}\left( {\frac{n}{f_{s}} - \tau_{bpf}} \right)}} \right)}} \right\rbrack}.$A low-pass filter 304 may be applied to the intermediate signal toextract its amplitude-modulating component

$\frac{a_{n}^{2}}{2}.$This component may be subtracted from the intermediate signal at adder318 to produce the signal

$\frac{a_{n}^{2}}{2}{\cos\left\lbrack {2\;\pi\;{f_{38\mspace{11mu}{kHz}}\left( {\frac{n}{f_{s}} - \tau_{bpf}} \right)}} \right\rbrack}$which is subsequently multiplied with an inverted version of theamplitude-modulating component at multiplier 320 in order to normalizethe signal. Hence a normalized 38 kHz sub-carrier signal

$\cos\left\lbrack {2\pi\;{f_{38{k{Hz}}}\left( {\frac{n}{f_{s}} - \tau_{bpf}} \right)}} \right\rbrack$is generated from digital implementation 300 of FDSN-COS block 104.

According to another aspect of the invention, a quadrature phaselook-ahead function block, such as block 106 in FIG. 1, may be providedfor introducing a phase delay to a 38 kHz sub-carrier signal fromFDSN-COS block 104, where the phase delay includes a compensation phaseφ₁ and a quadrature phase

$\frac{\pi}{2}.$The look-ahead scheme consequently produces a modified 38 kHzsub-carrier signal

${\cos\left( {{2\pi\;{f_{38{k{Hz}}}\left( {t - \tau_{bpf}} \right)}} + \phi_{1} + \frac{\pi}{2}} \right)}.$The compensation phase φ₁ of the modified signal should preferably besuch that it approximately cancels the non-ideal phase offset2πf_(38kHz)τ_(bpf). Therefore, quadrature look-ahead block 106 maysupply the normalized 38 kHz sub-carrier output signal from FDSN-COSblock 104 with a desired quadrature phase offset in place of the unknownfilter delay τ_(bpf).

FIG. 4 illustrates the steps of applying the signal recovery schemedescribed in FIG. 1 to an exemplary MPX signal having an ideal cosinepilot tone signal plot 500 a. After the recovered pilot signal isextracted from the MPX signal using a 19 kHz band-pass filter, such asfilter 102 in FIG. 1, a non-ideal delay (e.g., a 10⁻⁵ sec delay) isintroduced to the recovered pilot signal 500 as shown in plot 500 b. Inplot 500 b it can be observed that the delay also appears in anormalized 38 kHz signal 504 output from FDSN-COS functional block 104.Plot 500 d shows a recovered 38 kHz sub-carrier signal 506 after beingcorrected by quadrature phase look-ahead block 106. The resultingnormalized 38 kHz sub-carrier waveform 506 is offset by an approximatelyquadrature phase from the pilot tone signal 502 while propagating atabout twice the pilot frequency.

FIG. 5 depicts another illustrative embodiment of a 38 kHz sub-carriersignal recovery implementation 600 configured for cosine signalprocessing. According to this embodiment, a recovered pilot signala(t)cos(2πf_(19kHz)(t−τ_(bpf))) may be first extracted from an incomingMPX signal using a sharp band-pass filter 602 centered at 19 kHz. Thisrecovered pilot signal may be subsequently passed through

$a\frac{\pi}{4}$phase look-ahead functional block 604 which cancels the τ_(bpf) phasedelay in order to introduce an overall

$\frac{\pi}{4}$phase delay to the recovered pilot signal, hence generating aphase-shifted recovered pilot signal

${a(t)}{{\cos\left( {{2\pi\; f_{19{k{Hz}}}t} + \frac{\pi}{4}} \right)}.}$This

$\frac{\pi}{4}$look-ahead is ultimately used to produce a quadrature phase offset in a38 KHz sub-carrier output from FDSN-COS block 606. The resulting signalis again a standard sub-carrier waveform

${\cos\left( {{2\pi\; f_{38{k{Hz}}}t} + \frac{\pi}{2}} \right)},$where the requisite phase offset may be obtained by squaringphase-shifted recovered pilot signal of

${a(t)}{\cos\left( {{2\pi\; f_{19{k{Hz}}}t} + \frac{\pi}{4}} \right)}$in FDSN-COS block 606. Thus, instead of using a

$\frac{\pi}{2}$phase look-ahead block after FDSN-COS block 104 as seen in system 100 ofFIG. 1, a

$\frac{\pi}{4}$phase look-ahead block may be used before a FDSN-COS block as seen insystem 600 of FIG. 5.

According to another aspect of the invention, an exemplaryimplementation of

$\frac{\pi}{4}$phase look-ahead functional block 604 of FIG. 5 may be provided toimpose an overall

$\frac{\pi}{4}$phase delay in an input pilot signal, where the pilot signal issusceptible to filter noise. This look-ahead scheme operates byintroducing a compensation phase φ₂, combined with a

$\frac{\pi}{4}$offset, to the recovered pilot signal a(t)cos(2πf_(19kHz)(t−τ_(bpf))),yielding a phase-shifted pilot signal

${a(t)}{{\cos\left( {{2\pi\;{f_{19{k{Hz}}}\left( {t - \tau_{bpf}} \right)}} + \phi_{2} + \frac{\pi}{4}} \right)}.}$The scheme introduces an overall

$\frac{\pi}{4}$delay in the compensated signal by imposing the compensation phase φ₂ toapproximately cancel the unwanted phase delay 2πf_(19kHz)τ_(bpf).Accordingly, the compensation phase is mathematically expressed asφ₂=2πf_(19kHz)τ_(bpf). The signal

${a(t)}{\cos\left( {{2\pi\; f_{19{kHz}}^{t}} + \frac{\pi}{4}} \right)}$from phase look-ahead block 604 may subsequently be squared in FDSN-COSblock 606 to produce a quadrature phase offset in the resultingwaveform.

FIG. 6 illustrates the steps of applying the sub-carrier signal recoveryscheme described in FIG. 5 to an exemplary MPX signal that includes acosine pilot tone signal 702 as depicted in plot 700 a. A non-idealdelay (e.g., 10⁻⁵ sec delay) may be introduced to a filtered version 704of the recovered pilot signal as illustrated in plot 700 b. Plot 700 cshows a recovered pilot signal after being corrected by

$\frac{\pi}{4}$phase look-ahead block 604 described above with respect to FIG. 5. Theresulting signal is a phase compensated 19 kHz signal 706 with anapproximately

$\frac{\pi}{4}$phase delay from the ideal pilot tone signal 702. Plot 700 d illustratesa recovered 38 kHz sub-carrier signal 710 from the output of FDSN-COSfunctional block 606. This signal is offset by about a quadrature phasefrom the ideal pilot tone signal 702 while propagating at twice thepilot tone frequency.

FIG. 7 illustrates yet another illustrative embodiment of a 38 kHzsub-carrier signal recovery implementation 900 configured to processsignal that are represented as sine functions. The implementationincludes a 19 kHz band-pass filter 902 for recovering a pilot signalfrom an incoming MPX signal and a Frequency Doubling Scale Normalization(FDSN-SIN) functional block 904 for generating a sine 38 kHz sub-carriersignal in phase with the recovered pilot signal.

In FIG. 7, the received MPX signal may be expressed as a compilation ofthe following sine functions:MPX=0.9*M′+0.9*S′*sin(2πf _(38kHz) t)+(0.09±0.01)*sin(2πf _(19kHz)t),  Equation (2)where the first term corresponds to a baseband summation componentsignal, the second term corresponds to a 38 kHz-modulated differencecomponent signal, and the third term corresponds to a 19 kHz referencepilot tone signal. Both the summation and difference components may beexpressed in terms of L and R signals. In order to maximize a separationbetween the L and R signals for minimizing channel leakage, the sinesub-carrier signal of the second term in Equation (2) must preferably bein phase with the sine pilot tone signal of the third term in order tosatisfy the positive-slope requirement described above. This is incontrast to the cosine representation previously described wherein thereis preferably a quadrature phase difference between the 19 kHz pilottone signal and the 38 kHz sub-carrier signal.

To recover a 38 kHz sub-carrier signal from the scheme depicted in FIG.7, a sharp band-pass filter centered at 19 kHz may be used to recover apilot signal a(t)sin(2πf_(19kHz)t) from the MPX signal described inEquation (2), where a(t) is a slow-varying amplitude scaling function.

The sine recovered pilot signal may subsequently be applied to FDSN-SINfunctional block 1000 as depicted in detail in FIG. 8. First, anapproximated derivative of the recovered pilot signal2πf_(19kHz)a(t)cos(2πf_(19kHz)t) may be obtained at differentialoperation 1002. This estimated derivative signal may then be multipliedwith the original pilot signal at multiplier 1004 and based on thetrigonometric identity that sin 2θ=2 sin θ cos θ and an intermediatefrequency-doubled signal πf_(19kHz)a²(t)sin(2πf_(38kHz)t) is provided.The intermediate signal may then be normalized by a scaling function

$\frac{1}{\pi\; f_{19{kHz}}^{t}{a^{2}(t)}}$at scale normalization block 1006 to yield a sub-carrier signalsin(2πf_(38kHz)t).

FIG. 9 provides an exemplary digital-domain implementation 1100 ofFDSN-SIN functional block 1000. In particular, an approximate derivative

$2\kappa\; a_{n}{\sin\left( {2\pi\;{nf}_{19{kHz}}\frac{1}{f_{s}}} \right)}$of an recovered digital pilot signal

$a_{n}{\sin\left( {2\pi\;{nf}_{19{kHz}}\frac{1}{f_{s}}} \right)}$may be obtained at differential operation 1112. This approximatedderivative may then be multiplied with the recovered pilot signal atmultiplier 1114 to generate 38 kHz intermediate signal

$\kappa\; a_{n}^{2}{\sin\left( {2\pi\;{nf}_{38{kHz}}\frac{1}{f_{s}}} \right)}$from which its amplitude scaling factor ka_(n) ² may be extracted atoperations 1116-1126. The scaling factor may subsequently be multipliedwith the intermediate signal at multiplier 1128 to yield a normalized 38kHz sub-carrier signal

$\sin\left( {2\pi\;{nf}_{38{kHz}}\frac{1}{f_{s}}} \right)$in digital domain.

What has been described thus far are systems and methods for recoveringa 38 kHz sub-carrier signal from a 19 kHz pilot tone signal and forenforcing a pre-determined phase relation between the two signals toensure optimized channel separation. In particular, recovery schemes forboth cosine and sine signal are provided. In the case of recovering acosine sub-carrier signal from a recovered cosine pilot signal, twoexemplary implementations including respective phase look-ahead schemesare given. The phase look-ahead schemes are tailored to the respectivesignal recovery implementations such that a desired quadrature phaseoffset may be maintained in the recovered sub-carrier signal. Anexemplary implementation is also given for the recovery of a sinesub-carrier signal from a recovered sine pilot signal.

According to yet another aspect of the present invention, an exemplarydigital-domain implementation of the phase look-ahead schemes in FIGS. 1and 5 are provided. In particular, FIG. 10 depicts an illustrativeapproach to implement a

$\frac{\pi}{4}$look-ahead scheme in digital domain. Accordingly, one can approximate a

$\frac{\pi}{4}$look-ahead value y at any desired frequency f_(s) by applying a secondorder interpolation algorithm to an 19 kHz recovered pilot signal 802,where recovered pilot signal 802 may be delayed by τ_(bpf). Theinterpolation may be performed based on three sample points taken alongrecovered pilot signal 802 in a neighborhood to t, where t is a

$\frac{\pi}{4}$look-ahead frequency of f_(s) that also accounts for the recovered pilotsignal filter delay τ_(bpf). In one embodiment, the value of t may bedetermined according to the following equation:

$\begin{matrix}{t = {\left\lbrack {\frac{\frac{f_{s}}{19{kHz}}}{8} - 1} \right\rbrack + {\tau_{bpf}.}}} & {{Equation}\mspace{14mu}(3)}\end{matrix}$Once the look-ahead frequency t is determined, the signal value y at tmay be estimated using a second-order interpolation based on at leastthree sample points x⁻¹, x₀ and x₁ taken along recovered pilot signal802 at pre-defined distances away from t. In one embodiment, Look-aheadvalue y may be computed using the following interpolation scheme:

$\begin{matrix}{y = {{\frac{t\left( {t + 1} \right)}{2}x_{1}} + {\left( {1 - t^{2}} \right)x_{0}} + {\frac{t\left( {t - 1} \right)}{2}{x_{- 1}.}}}} & {{Equation}\mspace{14mu}(4)}\end{matrix}$

The quadrature look-ahead scheme of FIG. 1 may also be implemented indigital domain using a similar approach as described above based on aquadrature difference instead of a

$\frac{\pi}{4}$difference.

It should be understood that any other suitable approximation techniquesor variations on the second-order interpolation algorithm may be used inaccordance with the present invention. For example, if fastercomputation is desired, a linear interpolation algorithm may be used toestimate a signal value y based on two sample points in the neighborhoodof the look-ahead frequency t. If higher interpolation accuracy isdesired, higher order interpolations may be used to estimate alook-ahead value y based on four or more sample points in theneighborhood of t. Moreover, instead of interpolating sample points withlinear or polynomial functions, other forms of interpolation may beconstructed by choosing different classes of interpolants. In oneembodiment, rational interpolation may be used to perform interpolationby rational functions. In another embodiment, trigonometricinterpolation may be used to perform interpolation by trigonometricpolynomials that may include discrete Fourier transforms. In yet anotherembodiment, wavelets, or fast decaying oscillating waveforms, may beused to interpolate signals.

The illustrated embodiments of the invention are exemplary and do nolimit the scope of the invention. The equations described herein asbeing implemented by various blocks in the disclosed communicationsystem can be computed by hardware circuits and/or by softwareinstructions running on a processor. The equation computations need notbe performed with the exact terms and operations in the equations. Forexample, the equation computations can be performed using other termsand operations not shown in the equations to approximate the result ofcomputing the equations. Thus, the various blocks in the communicationsystem can perform computations based on the equations without directlycomputing the equations.

Additionally, the equations are exemplary and do not limit the scope ofthe invention. MPX signals may be described by equations other thenEquations 1 and 2. For example, a MPX signal may include a combinationof sine and cosine waveforms. Instead of using the second-orderinterpolation algorithm described in Equations 3 and 4 to implement thelook-ahead schemes of the present invention, other suitable estimationmethods may be used as described above. Further, while typical MPXcosine and sub-cosine frequencies are used, it should be understood thatany other suitable frequencies may also be used in accordance with theinvention.

Referring now to FIGS. 11A-11E, various exemplary implementations of thepresent invention are shown.

Referring now to FIG. 11A, the present invention can be implemented in ahigh definition television (HDTV) 1020. The present invention mayimplement either or both signal processing and/or control circuits,which are generally identified in FIG. 11A at 1022, a WLAN interfaceand/or mass data storage of the HDTV 1020. The HDTV 1020 receives HDTVinput signals in either a wired or wireless format and generates HDTVoutput signals for a display 1026. In some implementations, signalprocessing circuit and/or control circuit 1022 and/or other circuits(not shown) of the HDTV 1020 may process data, perform coding and/orencryption, perform calculations, format data and/or perform any othertype of HDTV processing that may be required.

The HDTV 1020 may communicate with mass data storage 1027 that storesdata in a nonvolatile manner such as optical and/or magnetic storagedevices. The HDD may be a mini HDD that includes one or more plattershaving a diameter that is smaller than approximately 1.8″. The HDTV 1020may be connected to memory 1028 such as RAM, ROM, low latencynonvolatile memory such as flash memory and/or other suitable electronicdata storage. The HDTV 1020 also may support connections with a WLAN viaa WLAN network interface 1029.

Referring now to FIG. 11B, the present invention implements a controlsystem of a vehicle 1030, a WLAN interface and/or mass data storage ofthe vehicle control system. In some implementations, the presentinvention may implement a powertrain control system 1032 that receivesinputs from one or more sensors such as temperature sensors, pressuresensors, rotational sensors, airflow sensors and/or any other suitablesensors and/or that generates one or more output control signals such asengine operating parameters, transmission operating parameters, and/orother control signals.

The present invention may also be implemented in other control systems1040 of the vehicle 1030. The control system 1040 may likewise receivesignals from input sensors 1042 and/or output control signals to one ormore output devices 1044. In some implementations, the control system1040 may be part of an anti-lock braking system (ABS), a navigationsystem, a telematics system, a vehicle telematics system, a lanedeparture system, an adaptive cruise control system, a vehicleentertainment system such as a stereo, DVD, compact disc and the like.Still other implementations are contemplated.

The powertrain control system 1032 may communicate with mass datastorage 1046 that stores data in a nonvolatile manner. The mass datastorage 1046 may include optical and/or magnetic storage devices forexample hard disk drives HDD and/or DVDs. The HDD may be a mini HDD thatincludes one or more platters having a diameter that is smaller thanapproximately 1.8″. The powertrain control system 1032 may be connectedto memory 1047 such as RAM, ROM, low latency nonvolatile memory such asflash memory and/or other suitable electronic data storage. Thepowertrain control system 1032 also may support connections with a WLANvia a WLAN network interface 1048. The control system 1040 may alsoinclude mass data storage, memory and/or a WLAN interface (all notshown).

Referring now to FIG. 11C, the present invention can be implemented in acellular phone 1050 that may include a cellular antenna 1051. Thepresent invention may implement either or both signal processing and/orcontrol circuits, which are generally identified in FIG. 11C at 1052, aWLAN interface and/or mass data storage of the cellular phone 1050. Insome implementations, the cellular phone 1050 includes a microphone1056, an audio output 1058 such as a speaker and/or audio output jack, adisplay 1060 and/or an input device 1062 such as a keypad, pointingdevice, voice actuation and/or other input device. The signal processingand/or control circuits 1052 and/or other circuits (not shown) in thecellular phone 1050 may process data, perform coding and/or encryption,perform calculations, format data and/or perform other cellular phonefunctions.

The cellular phone 1050 may communicate with mass data storage 1064 thatstores data in a nonvolatile manner such as optical and/or magneticstorage devices for example hard disk drives HDD and/or DVDs. The HDDmay be a mini HDD that includes one or more platters having a diameterthat is smaller than approximately 1.8″. The cellular phone 1050 may beconnected to memory 1066 such as RAM, ROM, low latency nonvolatilememory such as flash memory and/or other suitable electronic datastorage. The cellular phone 1050 also may support connections with aWLAN via a WLAN network interface 1068.

Referring now to FIG. 11D, the present invention can be implemented in aset top box 1080. The present invention may implement either or bothsignal processing and/or control circuits, which are generallyidentified in FIG. 11D at 1084, a WLAN interface and/or mass datastorage of the set top box 1080. The set top box 1080 receives signalsfrom a source such as a broadband source and outputs standard and/orhigh definition audio/video signals suitable for a display 1088 such asa television and/or monitor and/or other video and/or audio outputdevices. The signal processing and/or control circuits 1084 and/or othercircuits (not shown) of the set top box 1080 may process data, performcoding and/or encryption, perform calculations, format data and/orperform any other set top box function.

The set top box 1080 may communicate with mass data storage 1090 thatstores data in a nonvolatile manner. The mass data storage 1090 mayinclude optical and/or magnetic storage devices for example hard diskdrives HDD and/or DVDs. The HDD may be a mini HDD that includes one ormore platters having a diameter that is smaller than approximately 1.8″.The set top box 1080 may be connected to memory 1094 such as RAM, ROM,low latency nonvolatile memory such as flash memory and/or othersuitable electronic data storage. The set top box 1080 also may supportconnections with a WLAN via a WLAN network interface 1096.

Referring now to FIG. 11E, the present invention can be implemented in amedia player 1100. The present invention may implement either or bothsignal processing and/or control circuits, which are generallyidentified in FIG. 11E at 1104, a WLAN interface and/or mass datastorage of the media player 1100. In some implementations, the mediaplayer 1100 includes a display 1107 and/or a user input 1108 such as akeypad, touchpad and the like. In some implementations, the media player1100 may employ a graphical user interface (GUI) that typically employsmenus, drop down menus, icons and/or a point-and-click interface via thedisplay 1107 and/or user input 1108. The media player 1100 furtherincludes an audio output 1109 such as a speaker and/or audio outputjack. The signal processing and/or control circuits 1104 and/or othercircuits (not shown) of the media player 1100 may process data, performcoding and/or encryption, perform calculations, format data and/orperform any other media player function.

The media player 1100 may communicate with mass data storage 1110 thatstores data such as compressed audio and/or video content in anonvolatile manner. In some implementations, the compressed audio filesinclude files that are compliant with MP3 format or other suitablecompressed audio and/or video formats. The mass data storage may includeoptical and/or magnetic storage devices for example hard disk drives HDDand/or DVDs. The HDD may be a mini HDD that includes one or moreplatters having a diameter that is smaller than approximately 1.8″. Themedia player 1100 may be connected to memory 1114 such as RAM, ROM, lowlatency nonvolatile memory such as flash memory and/or other suitableelectronic data storage. The media player 1100 also may supportconnections with a WLAN via a WLAN network interface 1116. Still otherimplementations in addition to those described above are contemplated.

Thus it is seen that systems and methods are provided for efficient andaccurate recovery of a 38 kHz sub-carrier signal in an analog FMreceiver. One skilled in the art will appreciate that the invention canbe practiced by other than the described embodiments, which arepresented for purposes of illustration and not of limitation, and thepresent invention is limited only by the claims which follow.

1. A method for recovering a sub-carrier signal from a multiplexedsignal having an embedded pilot tone signal, the method comprising:receiving the multiplexed signal; recovering, with receiver circuitry, apilot signal from the received multiplexed signal, wherein the recoveredpilot signal includes a first phase offset from the embedded pilot tonesignal; generating a frequency-doubled signal from the recovered pilotsignal; and determining a second phase offset based on the first phaseoffset and a pre-determined phase offset; phase-shifting thefrequency-doubled signal by the second phase offset; and recovering thesub-carrier signal from the multiplexed signal based on the recoveredpilot signal and the phase-shifted frequency-doubled signal.
 2. Themethod of claim 1, wherein generating the frequency-doubled signal fromthe recovered pilot signal comprises: squaring the recovered pilotsignal to generate an intermediate signal; extracting a high-frequencycomponent of the intermediate signal; providing an amplitude scalingfunction; and multiplying the amplitude scaling function with thehigh-frequency component to generate the frequency-doubled signal. 3.The method of claim 2, wherein providing the amplitude scaling functioncomprises: delaying the intermediate signal to generate a delayedsignal; subtracting the high-frequency component from the delayed signalto generate a modulating function; and inverting the modulating functionto generate the amplitude scaling function.
 4. The method of claim 1,wherein generating the frequency-doubled signal from the recovered pilotsignal comprises: providing an estimated derivative of the recoveredpilot signal; multiplying the recovered pilot signal with the estimatedderivative to generate an intermediate signal; and multiplying theintermediate signal with an amplitude scaling function to generate thefrequency-doubled signal.
 5. The method of claim 1, wherein thepre-determined phase offset is approximately a quadrature phase offset.6. The method of claim 1, wherein the pre-determined phase offset isapproximately zero phase offset.
 7. A method for recovering asub-carrier signal from a multiplexed signal having an embedded pilottone signal, the method comprising: receiving the multiplexed signal;recovering, with receiver circuitry, a pilot signal from the receivedmultiplexed signal, wherein the recovered pilot signal includes a firstphase offset from the embedded pilot tone signal; determining a secondphase offset based on the first phase offset and a pre-determined phaseoffset; phase-shifting the recovered pilot signal by the second phaseoffset; generating a frequency-doubled signal from the phase-shiftedsignal; and recovering the sub-carrier signal from the multiplexedsignal based on the recovered pilot signal and the frequency-doubledsignal.
 8. The method of claim 7, wherein generating thefrequency-doubled signal from the phase-shifted signal comprises:squaring the phase-shifted signal to generate an intermediate signal;extracting a high-frequency component of the intermediate signal;providing an amplitude scaling function; and multiplying the amplitudescaling function with the high-frequency component to generate thefrequency-doubled signal.
 9. The method of claim 8, wherein providingthe amplitude scaling function comprises: delaying the intermediatesignal to generate a delayed signal; subtracting the high-frequencycomponent from the delayed signal to generate a modulating function; andinverting the modulating function to generate the amplitude scalingfunction.
 10. The method of claim 7, wherein the pre-determined phaseoffset is approximately a quadrature phase offset.
 11. A receiver systemfor recovering a sub-carrier signal from a multiplexed signal having anembedded pilot tone signal, the receiver system comprising: a receiverconfigured to receive the multiplexed signal; band-pass filter circuitryconfigured to recover a pilot signal from the received multiplexedsignal, wherein the recovered pilot signal includes a first phase offsetfrom the embedded pilot tone signal; frequency doubling scalenormalization circuitry configured to generate a frequency-doubledsignal from the recovered pilot signal; circuitry configured todetermine a second phase offset based on the first phase offset and apre-determined phase offset; phase-look ahead circuitry configured tophase-shift the frequency-doubled signal by the second phase offset; andreceiver circuitry configured to recover the sub-carrier signal from themultiplexed signal based on the recovered pilot signal and thephase-shifted frequency-doubled signal.
 12. The system of claim 11,wherein frequency doubling scale normalization circuitry configured togenerate the frequency-doubled signal from the recovered pilot signalcomprises: circuitry configured to square the recovered pilot signal togenerate an intermediate signal; circuitry configured to extract ahigh-frequency component of the intermediate signal; circuitryconfigured to provide an amplitude scaling function; and circuitryconfigured to multiply the amplitude scaling function with thehigh-frequency component to generate the frequency-doubled signal. 13.The system of claim 12, wherein the circuitry configured to provide theamplitude scaling function comprises: circuitry configured to delay theintermediate signal to generate a delayed signal; circuitry configuredto subtract the high-frequency component from the delayed signal togenerate a modulating function; and circuitry configured to invert themodulating function to generate the amplitude scaling function.
 14. Thesystem of claim 11, wherein frequency doubling scale normalizationcircuitry configured to generate the frequency-doubled signal from therecovered pilot signal comprises: circuitry configured to provide anestimated derivative of the recovered pilot signal; circuitry configuredto multiply the recovered pilot signal with the estimated derivative togenerate an intermediate signal; and circuitry configured to multiplythe intermediate signal with an amplitude-scaling function to generatethe frequency-doubled signal.
 15. The system of claim 11, wherein thepre-determined phase offset is approximately a quadrature phase offset.16. The system of claim 11, wherein the pre-determined phase offset isapproximately zero phase offset.
 17. A receiver system for recovering asub-carrier signal from a multiplexed signal having an embedded pilottone signal, the system comprising: a receiver configured to receive themultiplexed signal; band-pass filter circuitry configured to recover apilot signal from the received multiplexed signal, wherein the recoveredpilot signal includes a first phase offset from the embedded pilot tonesignal; circuitry configured to determine a second phase offset based onthe first phase offset and a pre-determined phase offset; phaselook-ahead circuitry configured to phase-shift the pilot signal by thesecond phase offset; frequency doubling scale normalization circuitryconfigured to generate a frequency-doubled signal from the phase-shiftedsignal; and receiver circuitry configured to recover the sub-carriersignal from the multiplexed signal based on the recovered pilot signaland the frequency-doubled signal.
 18. The system claim 17, whereinfrequency doubling scale normalization circuitry configured to generatethe frequency-doubled signal from the phase-shifted signal comprises:circuitry configured to square the phase-shifted signal to generate anintermediate signal; circuitry configured to extract a high-frequencycomponent of the intermediate signal; circuitry configured to provide anamplitude scaling function; and circuitry configured to multiply theamplitude scaling function with the high-frequency component to generatethe frequency-doubled signal.
 19. The system of claim 18, whereincircuitry configured to provide the amplitude scaling functioncomprises: circuitry configured to delay the intermediate signal togenerate a delayed signal; circuitry configured to subtract thehigh-frequency component from the delayed signal to generate amodulating function; and circuitry configured to invert the modulatingfunction to generate the amplitude scaling function.
 20. The system ofclaim 17, wherein the pre-determined phase offset is approximately aquadrature phase offset.